Beam former using phase difference enhancement

ABSTRACT

Noise discrimination in signals from a plurality of sensors is conducted by enhancing the phase difference in the signals such that off-axis pick-up is suppressed while on-axis pick-up is enhanced. Alternatively, attenuation/expansion are applied to the signals in a phase difference dependent manner, consistent with suppression of off-axis pick-up and on-axis enhancement. Nulls between sensitivity lobes are widened, effectively narrowing the sensitivity lobes and improving directionality and noise discrimination.

FIELD OF THE INVENTION

The invention relates to noise discrimination in signal detection andprocessing.

DESCRIPTION OF THE RELATED ART

FIG. 1 is a block diagram of a conventional real-time frequency domainsignal processing system 10 employing what is sometimes referred to asthe frequency sub-band method or the frame-overlap-and-add method. Thismethod uses a circuit 11 to divide incoming sampled temporal signalinformation into blocks of data referred to as frames. The sampled datacan be provided directly from a digital sensor or other processingsystem, or can be provided from an analog sensor or processing systemvia a standard Analog-to-Digital conversion (A/D or ADC) method (notshown). The frames can be adjacent or overlapping. Since the data aresamples of time domain data, all samples within a frame have noimaginary component, and the data is strictly “real.” If required by theapplication, these frames of data then may be multiplied in amultiplication circuit 12 by an analysis window 14 a to reduce artifactsthat can be introduced by subsequent transformation of the sampled timedata into the frequency domain. Subsequently, the windowed frames aretransformed to the frequency domain by any one of the many suchtransformations known to those of ordinary skill in the art, such as forexample the Hartley transform, the Wavelet transform, or the like. Themost commonly used of these transformations is the Fourier transform.Since the data is sampled and digitized, the DFT, or Discrete FourierTransform, is used in these cases, with a preference for using one ofthe fast-to-compute versions of this transform, known as the FastFourier Transform or FFT, represented at circuit 16.

Although there are choices for the analysis window, such as the Hanningwindow, that will reconstruct the time domain signal accurately withoutthe added complexity and computational cost of a synthesis window, suchanalysis windows suffer from accuracy compromises to achieve theimproved efficiency. Generally, a separate synthesis window 14 b isapplied by multiplication before the signal is reconstructed by theoverlap and add circuit, 19 (as shown in FIG. 1) to overcome thesecompromises, but at added cost.

Once in the frequency domain, the data is represented by complex numberscontaining both a “real” and an “imaginary” component. These complexnumbers, one for each frequency “bin” of the transform, represent themagnitude and relative phase angle of the temporal input signal dataaveraged over the time interval contained within the length of the frame(and weighted by the windowing function) as well as over the range offrequencies contained within the bandwidth of the “bin.” It is thisinput transform data that is then processed at circuit 17 by a selectedprocess to create an output transform of processed frequency domaindata.

Once the data is processed, the standard frequency domain method thencalls for inverse transformation of each frame of processed data tocreate a string of processed time domain frames of “real” data. Circuit18, denoting an inverse fast Fourier transform (IFFT) process, performsthis objective. If a synthesis window 14 b is used, then it is appliedat circuit 13 by multiplication of the output frame of time domain datawith the selected synthesis window: otherwise the output frame of datafrom circuit 18 is passed directly to circuit 19. Alternatively, thefrequency domain representation of the synthesis window can be appliedto the output from the signal process 17 by convolving the output fromthe process with the transformed synthesis window before performing theinverse Fourier transform at circuit 18. The time domain frames aresubsequently re-assembled by circuit 19 by performing concatenating oroverlapping-and-adding of the frames of processed real-time data tocreate the final digitized and sampled temporal output signal waveformcontaining the processed signal information. Of course, this sampledsignal can be, and often is, converted into an analog signal by the useof a standard Digital-to-Analog conversion (D/A or DAC) method (notshown) so that the processed output signal can be used in myriadapplications, such as scientific measurement, telephony, entertainmentsystems, communication systems, and so on.

Alternatively, the process can be applied in the time domain, wherein,for example, the input signal, either analog or digitized, is passedthrough a bank of bandpass frequency discrimination filters (eitheranalog or digital as appropriate). The outputs of each of the frequencyfilters is subsequently processed, and the processed signals are thencombined to form a processed output signal by adding the processedsignals together.

FIG. 2(a) shows the elements of a conventional prior art beamformingsystem, where a sensor system 21 provides two or more input signals 22that are time-aligned for the signal of interest. For best performance,these sensor signals should have matched sensitivity for all signals.The input sensor signals 22 provide the input data for the vectorsumming beamforming process of the system, as shown at circuit 23.

Although the vector summing process 23 is often performed as a vectoraverage, a vector average is simply a vector sum divided by a scalarnumber, and will simply be referred to hereinafter as a vector sum.

Consider one of the simplest beamforming sensor systems, the two-elementbroadside array 30 shown in FIG. 3. The two sensor elements 32 and 34 ofthis array are located on the axis X. It is well known that such abeamforming system can be steered using conventional signal delaymethods. In particular, conventional beam steering is accomplished byvarying relative phases of the input signals in such away that theincoming signal pattern is reinforced in a desired direction andsuppressed in undesired directions. The phase change is equivalent to atime delay—that is, the phase change at each frequency is a fixedoffset, and the phase change over frequency is linear. However, forsimplicity here it is assumed that the signal source of interest lies onthe sensitivity axis I of the array—that is, that the two sensor signalsare appropriately time delayed so as to be time-aligned for the desiredsignal of interest. When the sensor elements 32 and 34 areomni-directional and spaced one-half wavelength apart (180 electricaldegrees), the two-element broadside beamforming system, as shown in FIG.2(a), outputs a signal that is directly proportional to the vector sumof the two sensor element signals. This output has a sensitivity beampattern resembling a FIG. 8—that is, one having two sensitivity lobes 35and 36 as shown in FIG. 3. These lobes are maximum in the on-axisdirection, but are zero at ±90° azimuth directions (in the directions ofaxis X). These are the directions at which the electrical phasedifference between the sensor's signals is ±180° and therefore where thesignals cancel when summed together. The resulting low sensitivityregions 37 and 38 are referred to as “nulls.”

To improve the directionality of a sensor system normally impliesnarrowing the width of the main lobe(s) of sensitivity, which in FIG. 3is either lobe 35 or 36 (or both). In a conventional beamforming system,narrowing of the main sensitivity lobe is accomplished by incorporatingadditional sensor elements to enlarge the array, thereby increasing theacceptance aperture that concomitantly reduces the beam width. However,there are costs to this approach, including the additional sensorelements and associated amplifiers and A/D converters (in a digitalsystem) or filters (in an analog system), the added computational costsfor processing all the sensor signals, the result that the beam patternbecomes complex with many added side lobes in which the sensitivity ofthe system to unwanted signal sources is relatively high (that is, thesystem has relatively low noise immunity), the large physical size ofthe sensor array, and non-uniform frequency response for off-axissignals, among others.

For these reasons, another method called “super resolution” beamforminghas been employed, wherein the increased aperture is filled withadditional sensor elements, but the elements are non-uniformly spacedand the resulting sensor signals are non-uniformly weighted inamplitude. In such a system (not shown), the width of the main lobe ofsensitivity can be more greatly narrowed as compared to a similarbeamforming system with uniformly spaced sensor elements. However, to besuccessful the super resolution approach still requires a great numberof sensor elements and associated circuitry and suffers fromsignificantly increased computational costs, high side lobe sensitivity,large physical size, and non-uniform off-axis frequency response.

In order to address the side lobe pickup problem, another method hasbeen employed in which additional beamformer systems are used with thesame set of array sensor signals. The additional beamformers createsensitivity beams that are in the directions of the side lobes of themain beamformer. The output signals from these additional beamformersare then scaled and subtracted from the output signal from the mainbeamformer in order to partially cancel the main beamformer's sidelobes. In general, although the side lobes can be reduced with such anapproach, the tradeoffs include a wider main lobe, high complexity andcost, and the retention of a high number of sensors.

Yet another category of conventional beamformer is the generalized sidelobe canceller (GSC) where a multiple sensor system is combined with anull-steering method. In this technology, the sensitivity toward thedesired source is maintained constant while one or more of the nulls aresteered toward detected off-axis noise sources. Examples of this type ofbeamforming system are the well known Griffiths-Jim beamformer and theFrost beamformer. In this type of beamforming system the number ofdiscrete noise sources that can be nulled is equal to the number ofindependently steerable nulls, and the number of independently steerablenulls is equal to one less than the number of sensors. Thus, to beeffective in most real-life situations where there are numerous noisesources and multiple-reflections of those noise sources, the number ofsensors must be large, along with the associated high system complexity,large compute power requirement, and high cost. Further, such systems,because the nulls are very narrow, require adaptive circuit techniquesto accurately center the nulls on the noise source directions, and theseadaptive methods are slow to adapt, allowing significant noise to passduring the adaptation time.

One common characteristic of these prior art systems is that the null ornulls created by these methods are quite narrow. As more sensor elementsare incorporated, more nulls are created and the numerous resultingnulls are narrower yet.

BRIEF SUMMARY OF THE INVENTION

In accordance with one aspect of the invention, there is provided amethod for improving noise discrimination in a system having a pluralityof sensors each generating a sensor input signal representable by aninput vector having phase and magnitude components in response to asignal stimulus, the plurality of sensors being arranged to have anon-axis direction. This method includes generating from at least twoinput vectors an input phase difference value, enhancing the input phasedifference value as a function of the location of the signal stimulusrelative to the on-axis direction, generating two output vectorscorresponding to the two input vectors, the two output vectors having aphase difference based on the enhanced input phase difference value, andcombining the two output vectors.

In accordance with a further aspect of the invention, there is provideda method for improving noise discrimination in a system having aplurality of sensors each generating a sensor input signal representableby an input vector having phase and magnitude components in response toa signal stimulus, the plurality of sensors being arranged to have anon-axis direction. This method includes generating an attenuation factoras a function of a phase difference from two input vectors, combiningthe two input vectors to obtain an output vector, and attenuating theoutput vector by the attenuation factor.

In accordance with a further aspect of the invention, there is provideda method for improving noise discrimination in a system having aplurality of sensors each generating an input signal representable by aninput vector having a phase component and a magnitude component, theplurality of sensors arranged to have an on-axis direction. The methodincludes using a first pair of sensors to obtain a coarse vector phasedifference corresponding to a coarse measurement of an angle of arrivalof a signal input source relative to the on-axis direction, using asecond pair of sensors to obtain a fine vector phase differencecorresponding to a fine measurement of the angle of arrival of thesignal input source, generating an input phase difference value from thecoarse and fine vector phase differences, enhancing the input phasedifference value as a function of the angle of arrival to generate anoutput phase difference value, generating first and second outputvectors having a phase difference based on the output phase differencevalue, and combining the first and second output vectors.

In accordance with a further aspect of the invention, there is provideda method for improving noise discrimination in a system having aplurality of sensors each generating an input signal representable by aninput vector having a phase component and a magnitude component, theplurality of sensors arranged to have an on-axis direction. This methodincludes using a first pair of sensors to obtain a coarse vector phasedifference corresponding to a coarse measurement of an angle of arrivalof a signal input source relative to the on-axis direction, using asecond pair of sensors to obtain a fine vector phase differencecorresponding to a fine measurement of the angle of arrival of thesignal input source, generating an attenuation factor as a function ofthe coarse and fine vector phase differences, combining the inputvectors corresponding the second pair of sensors to obtain an Outputvector, and attenuating the output vector by the attenuation factor.

In accordance with a further aspect of the invention, there is provideda method for enhancing regional sensitivity noise discrimination fromfirst and second pairs of sensors, each sensor generating a sensor inputsignal representable by an input vector having a magnitude and phase.This method includes applying a first process to the first pair ofsensors to obtain a first output corresponding to sensitivity in a firstregion, applying a second process to the second pair of sensors toobtain a second output corresponding to sensitivity in a second region,and combining the first and second outputs. The first process includesenhancing an input phase difference value corresponding to a phasedifference between signals from first and second sensors in the firstpair of sensors.

In accordance with a further aspect of the invention, there is provideda method for enhancing regional sensitivity noise discrimination fromfirst and second pairs of sensors, each sensor generating a sensor inputsignal representable by an input vector having a magnitude and phase.The method includes applying a first process to the first pair ofsensors to obtain a first output corresponding to sensitivity in a firstregion, applying a second process to the second pair of sensors toobtain a second output corresponding to sensitivity in a second region,and combining the first and second outputs. The first process includesattenuating an output vector obtained by combining first and secondinput vectors corresponding to signals from first and second sensors ofthe first pair of sensors by an attenuation factor which is a functionof a difference in phase between the first and second input vectors.

In accordance with a further aspect of the invention, there is provideda method for accommodating device and/or signal mismatch in a sensorarray system including first and second sensors generating first andsecond input signals representable at at least one frequency by firstand second input vectors each having a phase component and a magnitudecomponent. The method includes, at the at least one frequency, using themagnitude of the first and second input vectors to obtain correspondingfirst and second mathematically mean matched vectors.

In accordance with a further aspect of the invention, there is provideda beamformer using a plurality of sensors each producing a sensor inputsignal representable by an input vector having phase and magnitudecomponents. The beamformer includes a combining circuit for receivingthe sensor input signals and generating a combined signal therefrom, afirst differencing circuit for receiving the sensor input signals andgenerating a first difference signal therefrom, an adaptive filter forreceiving the difference signal and generating a filtered signaltherefrom, a second differencing circuit for receiving the filteredsignal and a delayed version of the combined signal and generating anoutput signal therefrom, and a phase difference enhancement circuit forenhancing a phase difference of input vectors representing sensor inputsignals from the plurality of sensors.

In accordance with a further aspect of the invention, there is provideda beamformer using a plurality of sensors each producing a sensor inputsignal representable by an input vector having phase and magnitudecomponents, the beamformer including a combining circuit for receivingthe sensor input signals and generating a combined signal therefrom afirst differencing circuit for receiving the sensor input signals andgenerating a first difference signal therefrom an adaptive filter forreceiving the difference signal and generating a filtered signaltherefrom, a second differencing circuit for receiving the filteredsignal and a delayed version of the combined signal and generating anoutput signal therefrom, and a phase difference responsive circuit forreceiving at least one of the signals, and modifying that signal toproduce a modified signal that is a function of the phase difference ofthe input signals from the plurality of sensors.

In accordance with a further aspect of the invention, there is provideda beamformer using a plurality of sensors each producing a sensor inputsignal representable by an input vector having phase and magnitudecomponents. The beamformer includes a processing circuit for receivingthe sensor input signals and generating a processed signal therefrom,the processing circuit including a first phase difference enhancementcircuit for enhancing the phase difference of input vectors representingsensor input signals from the plurality of sensors, a first differencingcircuit for receiving the sensor input signals and generating a firstdifference signal therefrom, an adaptive filter for receiving thedifference signal and generating a filtered signal therefrom, a seconddifferencing circuit for receiving the filtered signal and a delayedversion of the processed signal and generating an output signaltherefrom, and a second phase difference enhancement circuit forenhancing a phase difference of input vectors representing sensor inputsignals from the plurality of sensors.

In accordance with a further aspect of the invention, there is provideda method for time domain processing of signals from a plurality ofsensors. The method includes obtaining from the plurality of sensors aplurality of corresponding input signals, applying the input signals toa bank of bandpass frequency discrimination filters to thereby obtain afiltered signal from each filter, generating phase angle differencevalues from the filtered signals, attenuating each of the plurality ofinput signals by an attenuation factor which is a function of the phaseangle difference values, and combining the plurality of attenuated inputsignals.

In accordance with a further aspect of the invention, there is provideda method for time domain processing of signals from a plurality ofsensors, the method including obtaining from the plurality of sensors aplurality of corresponding input signals each representable by an inputvector having phase and magnitude components, applying the input signalsto a bank of bandpass frequency discrimination filters to thereby obtaina filtered signal corresponding to each sensor from each filter,generating, for each filter, an instantaneous phase angle differencevalue representative of the phase angle difference between the filteredsignals from that filter, enhancing the phase component of each filteredsignal by an enhancement value which is a function of the instantaneousphase angle difference value associated with that filter to therebyobtain enhanced output signals, and combining the enhanced outputsignals.

In accordance with a further aspect of the invention, there is provideda pickup device that includes at least first and second sensorsgenerating first and second sensor input signals, respectively, inresponse to a signal stimulus, the first and second input signals beingrepresentable by first and second input vectors each having a phasecomponent and a magnitude component. The pickup device also includes atleast one circuit adapted to: generate from the first and second senorinput vectors an input phase difference value; enhance the input phasedifference value as a function of the location of the signal stimulusrelative to an on-axis direction of the at least first and secondsensors; generate two output vectors corresponding the first and secondinput vectors, the two output vectors having a phase difference based onthe enhanced input phase difference value; and combine the two outputvectors.

In accordance with a further aspect of the invention, there is provideda system for improving noise discrimination in at least first and secondinput signals representable by first and second input vectors eachhaving a phase component and a magnitude component. The system includesa first circuit adapted to generate an attenuation factor as a functionof a phase difference of the first and second input vectors, a combinerfor combining the first and second input vectors into an output vector,and an attenuation circuit for attenuating the output vector by theattenuation factor.

In accordance with a further aspect of the invention, there is provideda device for improving noise discrimination. The device includes firstand second pairs of sensors arranged to have an on-axis direction, eachsensor generating an input signal representable by an input vectorhaving a phase component and a magnitude component. The device furtherincludes at least one circuit adapted to: generate from the first pairof sensors a coarse vector phase difference corresponding to a coarsemeasurement of an angle of arrival of a signal input source relative tothe on-axis direction; generate from the second pair of sensors a finevector phase difference corresponding to a fine measurement of the angleof arrival of the signal input source; generate an input phasedifference value from the coarse and fine vector phase differences;

enhance the input phase difference value as a function of the angle ofarrival to generate an output phase difference value; generate first andsecond output vectors having a phase difference based on the outputphase difference value; and combine the first and second output vectors.

In accordance with a further aspect of the invention, there is provideda device from improving noise discrimination, the device including firstand second pairs of sensors arranged to have an on-axis direction, eachsensor generating an input signal representable by an input vectorhaving a phase component and a magnitude component. The device alsoincludes at least one circuit adapted to: generate from the first pairof sensors a coarse vector phase difference corresponding to a coarsemeasurement of an angle of arrival of a signal input source relative tothe on-axis direction; generate from the second pair of sensors a finevector phase difference corresponding to a fine measurement of the angleof arrival of the signal input source; generate an attenuation factor asa function the coarse and fine vector phase differences; combine theinput vectors corresponding to the second pair of sensors to obtain anoutput vector; and attenuate the output vector by the attenuationfactor.

In accordance with a further aspect of the invention, there is provideda system exhibiting enhanced regional sensitivity noise discrimination.The system includes first and second pairs of sensors, each sensorgenerating a sensor input signal representable by an input vector havinga magnitude and phase, a at least one circuit adapted to: apply a firstprocess to the first pair of sensors to obtain a first outputcorresponding to sensitivity in a first region; apply a second processto the second pair of sensors to obtain a second output corresponding tosensitivity in a second region; and combine the first and secondoutputs. The first process includes enhancing an input phase differencevalue corresponding to a phase difference between signals from first andsecond sensors in the first pair of sensors.

In accordance with a further aspect of the invention, there is provideda system exhibiting enhanced regional sensitivity noise discrimination.The system includes first and second pairs of sensors, each sensorgenerating a sensor input signal representable by an input vector havinga magnitude and phase, and at least one circuit adapted to: apply afirst process to the first pair of sensors to obtain a first outputcorresponding to sensitivity in a first region; apply a second processto the second pair of sensors to obtain a second output corresponding tosensitivity in a second region; and combine the first and secondoutputs. The first process includes attenuating an output vectorobtained by combining first and second input vectors corresponding tosignals from first and second sensors of the first pair of sensors by anattenuation factor which is a function of a difference in phase betweenthe first and second input vectors.

In accordance with a further aspect of the invention, there is provideda sensitivity matching circuit adapted to accommodate device and/orsignal mismatch in a sensor array system that includes first and secondsensors generating first and second input signals representable at atleast one frequency by first and second input vectors each having aphase component and a magnitude component. The sensitivity matchingcircuit includes one or more circuits adapted to use the magnitude ofthe first and second input vectors to obtain corresponding first andsecond mathematically mean matched vectors.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

Many advantages of the present invention will be apparent to thoseskilled in the art with a reading of this specification in conjunctionwith the attached drawings, wherein like reference numerals are appliedto like elements, and wherein:

FIG. 1 is a block diagram of a conventional real-time frequency domainsignal processing system 10 employing what is sometimes referred to asthe frequency sub-band method or the frame-overlap-and-add method;

FIG. 2A is a block diagram showing the elements of a conventionalbeamforming system in which a sensor system 21 provides two or moreinput signals 22 that are time-aligned for the signal of interest;

FIG. 2B is a block diagram showing the elements of an inventive systemin which a phase enhancement process 24 is disposed between theacquisition of the sensor signals and the beamforming process;

FIG. 2C is a block diagram showing the inventive use of a signalsensitivity matching circuit located in advance of a phase enhancementcircuit depicted in FIG. 2;

FIG. 3 is a schematic diagram of a conventional beamforming sensorsystem consisting of a two-element broadside array;

FIG. 4 is a depiction of a principle behind one aspect of the invention,in which two sensor elements A and B are shown arranged along a line Xina broadside array configuration 40, and an off-axis noise source, N, isshown at the physical azimuthal arrival angle θ_(N) away from the axis Iof maximum sensitivity;

FIG. 5 is a flow diagram illustrating an implementation of one aspect ofthe invention;

FIGS. 6A and 6B are graphs depicting the effect of applying some of theinventive formulae to enhance the angular phase difference between twoinput signal vectors;

FIGS. 7A-7F are vector diagrams showing some of the inventive principlesinvolved in combining input signals;

FIG. 8A is a graphical illustration of the performance of a conventionalbeamforming system using two cardioid microphone sensor elements spaced7-cm apart;

FIG. 8B is a graphical illustration of the performance of a system inaccordance with the invention using the same microphone array as that ofFIG. 8A, and the expansion function given by Equation 1 with a sharpnessvalue S_(D) of 10 at 1000 Hz;

FIG. 8C is a graphical illustration showing the beam shape at 1000 Hzfor a conventional beamforming system in which additional elements havebeen added to make the main sensitivity lobe's FWHM (full width at halfmaximum—a standard method of measuring beam widths) equal to that of thenovel system under the same conditions as described for FIG. 8B,achieved using 13 sensor elements, all spaced 7-cm apart for a totalaperture (array) size of over 85-cm (assuming readily available 6-mmdiameter electret microphone elements);

FIG. 8D is a graphical illustration of an approach in accordance withthe invention providing the 1000 Hz beam pattern;

FIG. 8(E) shows the 1000 Hz beam pattern produced by the novel systemwhen the sharpness parameter S is increased to a value of 20.

FIG. 9 is a flow diagram of a signal sensitivity matching system asimplemented within the framework of a beamforming system in accordancewith the invention;

FIG. 10 is a vector diagram showing the input signal vectors A and Bforming an isosceles triangle when the signal magnitudes are matched;

FIG. 11 is a block diagram showing a more computationally efficientapproach utilizing a signal attenuation characteristic directly, insteadof first calculating the expanded phase vectors A′ and B′, in accordancewith the invention;

FIG. 12 is a flow diagram showing how the attenuation ratio can be usedto provide another way of implementing the inventive noise reductionmethod;

FIG. 13 is a graph of the attenuation value created, using Equation 4,and the phase enhancement function of Equation 1 in accordance with theinvention;

FIG. 14 is a flow diagram of a computationally efficient approach foraccomplishing noise reduction in accordance with the invention;

FIG. 15 is a graph of, and the defining equations for, some typicalattenuation functions that can be used with a beamforming noisereduction system in accordance with the invention;

FIG. 16 is a schematic diagram showing a method for both extending thenovel method to linear broadside arrays of greater than two elements, ameans for resolving the input signal electrical phase differenceambiguity created by greater sensor spacings;

FIGS. 17A and 17B are schematic diagrams of two approaches for producinga range sensitive beam pattern in accordance with the invention;

FIGS. 18A-18C are schematic diagrams of three different approaches forcreating a “pencil” beam—that is, a beam with both reduced azimuthal(width) and reduced elevational (height) extent—in accordance with theinvention;

FIG. 19A is a schematic diagram of a prior art two-element noisereduction system; and

FIGS. 19B and 19C are schematic diagrams showing the inventive use ofphase enhancement process in a Griffiths-Jim beamformer arrangement.

DETAILED DESCRIPTION OF THE INVENTION

In accordance with an aspect of the invention, a novel approach based onenhancing the performance of beamforming systems is disclosed. As ageneral aim, an aspect of the invention operates on the principle ofenhancing or enlarging the nulls of a beam pattern created by such abeamforming system.

The novel approach, in accordance with an aspect of the invention, is towiden the nulls—that is, regions 37 and 38 in FIG. 3—rather than tonarrow the main lobes 35 and 36 of a beamforming system. This approachimproves directionality, but by way of a unique and advantageousapparatus and method. By widening the nulls using the inventive method,the improved directionality is accomplished without increasing thenumber of sensor elements and associated amplifiers and A/D converters(in a digital system) or filters (in an analog system), with reducedcomputational costs for processing the sensor signals, with the resultthat the beam pattern is simple without added side lobes and theirincreased sensitivity to unwanted noise signal sources, with smallphysical size of the sensor array, with low system hardware costs,without long adaptation times, and with the added ability to produceuniform frequency response for off-axis signals, among other benefits.It will be appreciated that while for simplicity, the followingdescriptions will discuss a two-sensor implementation of the invention,the same techniques are extendable to arrays having a greater numberthan two, in one-, two-, and three-dimensional arrangements.

As shown in FIG. 2(b), in one aspect of the invention, a phaseenhancement process 24 is disposed between the acquisition of the sensorsignals 21 and the beamforming process 23. The phase enhancement process24 produces phase-enhanced signals 25 that are then used as inputsignals by the beamforming process 23.

A principle behind one aspect of the invention is illustrated in FIG. 4,showing two sensor elements A and B arranged along a line X in abroadside array configuration 40, and an off-axis noise source, N,located at the physical azimuthal arrival angle φ_(N) away from the axisI of maximum sensitivity. Because there are two input signals for thissystem, one from each of sensor elements A and B, two Fourier inputsignal transforms are available to the process. Each transform consistsof many frequency “bins” of data, and each data value in a bin is acomplex number Z, whereinZ=M cos θ+iM sin θcontaining information about both the magnitude (M) and the relativesignal phase (θ) of each signal during a particular interval oftime—that is, a particular frame.

Within a frame, for example for input signal A, the value in the nth binof its input Fourier transform is:Z _(A)(n)=M _(A)(n)cos θ_(A)(n)+iM _(A)(n)sin θ_(A)(n)where M_(A)(n) is the average magnitude of input signal A for thefrequencies represented by frequency bin n, and where θ_(A)(n) is theaverage relative signal phase of input signal A for the frequenciesrepresented by the same frequency bin n. The signal phase is oftenreferred to as the “electrical phase” of the signal.

Similarly, for input signal B, the value in the n^(th) bin of its inputFourier transform is:Z _(B)(n)=M _(B)(n)cos θ_(B)(n)+iM _(B)(n)sin θ_(B)(n)where M_(B)(n) is the average magnitude of input signal B for thefrequencies represented by frequency bin n, and where θ_(B)(n) is theaverage relative signal phase of input signal B for the frequenciesrepresented by frequency bin n. Thus, for each frequency correspondingto a bin there are available to the process two complex numbers allowingthe calculation of two relative input signal phase angle values, namelyθ_(A)(n) and θ_(B)(n).

Henceforth, for simplicity it will be assumed that each calculation isperformed on a bin-by-bin basis and the frequency bin index n will bedropped.

FIG. 5 is a flow diagram illustrating an implementation of one aspect ofthe invention. At 51 a and 51 b, magnitude and phase information of thesignals from sensors A and B is obtained. For each bin pair, thedifference between the two relative input signal electrical phase anglevalues is calculated at 52. In other words:Δθ₁=Δθ_(A)−Δθ_(B) or, alternatively Δθ₁=Δθ_(B)−Δθ_(A)where θ_(A) or θ_(B) is the arc tangent of the ratio of the imaginarypart of the input signal divided by the real part of part of the inputsignal , and Δθ_(I) is the signal electrical phase angle differencebetween the two input signals A and B for each frequency bin pair.

Although the mathematical method shown above is theoretically correct,in practical (real-world) systems, the arc tangent function usuallygenerates a relative phase value that is restricted to the interval−π≦Δθ<π. Thus, when calculating the input signal phase difference anglevalue Δθ_(I), the calculated result is on the interval −2π≦Δθ<2π.Although this value can be used directly to accomplish the inventiveprocess, for mathematical reasons it is often more convenient if thevalue lies on the interval −π≦Δθ<π. The calculated input signal phasedifference angle value Δθ_(I) can be “re-wrapped” to lie on the desiredinterval by the process of adding 2π when the value is less than −π, andsubtracting 2π when the value is more than π. No change is made when thevalue already lies on the interval −π≦Δθ<π. After this calculation, theresulting value for Δθ_(I) lies on the desired interval −π≦Δθ<π.

After re-wrapping the resulting phase difference value, an inputelectrical phase difference number representing the input signal phasedifference value between the two sensor signals is generated. In theory,since the signal of interest lies on the sensing axis I of the arraysystem 40, in other words since the portion of signals A and Brepresenting the desired signal are time-aligned, there will be no phasedifference for that signal and the phase difference number should bezero. However, for signals arriving from unwanted off-axis “noise”sources, N, there will be an electrical phase difference, and the phasedifference number is a function of the azimuthal angle of arrival,φ_(N).

With reference to FIG. 4, it will be appreciated that the followingapplies:Δθ_(I) =π·f{√{square root over (s²+4(D²+Ds·sin φ_(N)))}− √{square rootover (s²+4(D²−Ds·sin φ_(N)))}}/2cwhere f is the center frequency for the frequency bin, s is the physicalspacing between the sensor elements, D is the distance from the centerof the sensor array to the noise source N, c is the propagation speed ofthe signal (here it is the speed of sound in air), and φ_(N) is theazimuthal angle of arrival of the signal from noise source N.

If D>>s, in other words if the noise source N is located a significantdistance from the array, then the electrical phase difference numbersimplifies to:${\Delta\quad\theta_{I}} = {\pi\quad{\frac{f \cdot s}{c} \cdot \sin}\quad\phi_{N}}$

By making the system of the invention “think” that the arrival of mostoff-axis noise signals is from sources that are near ±90° azimuth, thesesignals are made to fall in the nulls, and are then considerablyattenuated by the subsequent beamforming process of signal vectorsummation. In accordance with one aspect of the invention, this isaccomplished by expanding the measured input electrical phase differencenumber Δθ_(I) toward 180° at 53 in FIG. 5 using an appropriate expansionfunction.

It will be appreciated that vector summation includes both summationwith and without first inverting the signals provided to the summationcircuit. In general, broadside array beamforming does not utilize signalinversion, while end-fire array beamforming does. Both types ofbeamforming systems are contemplated to be within the scope of thisinvention.

Additionally, for the purposes of this invention, phase enhancementincludes both phase expansion using an expansion function, as discussedabove, and phase compression, as will be described below. Phaseexpansion applies in many array systems, such as the broadsidebeamformer, for narrowing the main sensitivity lobe. Alternatively, insignal differencing array systems, such as are many end-fire arraysystems, phase compression is required to narrow the main sensitivitylobe. However, there are applications wherein instead the nulls are tobe narrowed, and in such systems phase expansion and phase compressionare also contemplated. This is discussed, for example, below inreference to the GSC beamforming system.

Now considering the case where summation is performed without inversion,many functions may be used for expanding the input electrical phasedifference number. In one embodiment of this invention, wherein thedesired source of acoustic signals is time-aligned such that theelectrical signals produced by the sensor elements are in phase, all ofthe available expansion functions will have one property in common: theywill not change a phase difference that is at 0°, since signals withsuch a difference is most likely from the desired source and should notbe attenuated. However, as the electrical phase difference between theinput signals increases (either plus or minus) away from 0°, there isincreasing likelihood that the signal pair originates from undesiredoff-axis noise sources. Thus, for example, an electrical phasedifference of 45° can be expanded to for example 80° before the twosignals are combined in the beamforming process. Such an expansion willdecrease the magnitude of the output signal, since the two signals aresummed in the beamforming process, and the two signals will be more outof phase after the expansion. As the input phase angle differenceincreases, the expanded output difference is moved more and more toward±180°. Thus, for example, an electrical phase difference of 90° can beexpanded to 179° before the two signals are combined in the beamformingprocess, giving a nearly complete attenuation for such signals.

Expanding the input electrical phase angle difference number Δθ_(I) tocreate the expanded output signal phase angle difference number Δθ_(O)is accomplished by applying an appropriate expansion function having thecharacteristics just described. One such function is: $\begin{matrix}{{\Delta\quad\theta_{O}} = {\pi \cdot {{sgn}( {\Delta\quad\theta_{I}} )} \cdot \{ {1 - \lbrack {1 - ( \frac{{\Delta\quad\theta_{I}}}{\pi} )} \rbrack^{S}} \}}} & (1)\end{matrix}$where the angles Δθ_(I) and Δθ_(O) are expressed in radians, and S is aparameter that controls the narrowness or sharpness of the resultingsensitivity beam, 1<S≦∞.

FIG. 6(a) is a graph depicting the effect of applying this formula toenhance the angular phase difference between the two input signalvectors. The input electrical phase difference number Δθ_(I) is plottedon the ordinate, or horizontal axis, while the expanded electrical phaseangle difference number Δθ_(O) is plotted on the abscissa, or verticalaxis.

When there is no expansion, for example when the sharpness parameter Sis set equal to 1 in the above equation, the output signal phase angledifference number Δθ_(O) is equal to the input signal phase angledifference number Δθ_(I)—that is, Δθ_(O)=Δθ_(I)—and the system operateslike a conventional beamforming system. This condition is shown by thediagonal graph curve 60 in FIG. 6(a).

However, for larger values of S, the phase difference is expanded and acommensurate improvement of the sensitivity beam pattern is realized.The graph curve 62 shown in FIG. 6(a) illustrates the expansion curvethat results from setting the sharpness parameter S equal to a value of10 in Equation 1. Note that the curve passes through the point 0,0 sothat no change is effected for signals arriving from the desired sourcelocation that is on the main sensitivity axis. For signals arriving fromazimuth angles away from the main sensitivity axis, the input electricalphase angle difference number Δθ_(I) has a non-zero value and theresulting enhanced output electrical phase angle difference numberΔθ_(O) is thereby changed away from the original input electrical phaseangle difference value and toward ±180 electrical degrees when expansionis effected commensurate with curves 62-64.

Curves 63 and 64 show the phase expansion characteristics of Equation 1for S values of 5 and 20 respectively. Thus, for this equation, as thesharpness value is increased, the phase enhancement increases. Thisprovides a method for setting the resulting beam width in a particularapplication to exclude the pickup of off-axis noise signals as requiredfor that application. However, it also provides a method of controllingthe resulting beam characteristics as a function of any other parameter,for example to frequency-compensate the system's sensitivity to off-axissignals by varying the value of the parameter S as a function offrequency in order to create a constant width beam across allfrequencies. Alternatively, the sharpness parameter S may be varied inreal time to provide beam control in real time.

It is instructive to compare at this juncture the behavior of aconventional beam steering system. It will be recalled that in such asystem, beam steering is accomplished by varying relative phases of theinput signals in such a way that the incoming signal pattern isreinforced in a desired direction and suppressed in undesireddirections. The phase change is equivalent to a time delay—that is, thephase change at each frequency is a fixed offset, and the phase changeover frequency is linear. Since unlike in the presently claimedinvention there is no phase enhancement in a conventional beam steeringsystem (that is, “S” in such a system, if expressed in the language ofthe invention, would have a value of 1), the conventional beam steeringsystem curve in FIG. 6 a would assume a straight line, parallel to line60, and not passing through the point 0,0. Such a line is designated at69 in FIG. 6 a.

FIG. 6(b) shows examples of additional enhancement functions and theresulting enhancement curve for each. As exemplified by the curveslabeled 65 and 66, the phase need not be expanded at every value awayfrom the point 0,0. For these curves, there is compression over alimited range near 0,0, but phase expansion occurs at input differencevalues further away. The curve labeled 67 demonstrates that expansionmay also be limited to input electrical phase difference numbers near tothe point 0,0, while no expansion or even compression may occur forinput phase difference values near ±180 degrees.

This discussion has addressed only a few examples of the possibleenhancement formulas and curves, and is not intended to be limiting.Formulae that include the point 0,0, and curves that pass through thepoint 0,0, and expand the phase difference—in other words increase thephase difference—at other points conform to one aspect of the invention.Formulas and curves that maintain a constant phase difference at someother selected point, and expand the phase difference at other pointsconform to another aspect of the invention. In accordance with anotheraspect of the invention, expansion is only applied to some input phaseangle difference number values, Δθ_(I). In practice, phase expansionwill likely be applied, to a greater or lesser extent, to most values,although it will be recognized that there is no requirement thatexpansion be applied to most or even a substantial portion of thevalues. Further, in some applications, the phase enhancement may beapplied using a look-up table of discrete values, rather than by using acontinuous function or curve, and while the term phase enhancement isused in a general sense, it will be recognized that compression, orreduction of the phase difference is included in the general concept ofphase enhancement as referred to herein.

As seen from Equation 1 above, the sign of the input phase angledifference number Δθ_(I) is used separately from its magnitude. Sincethe magnitude never takes on negative values, the magnitude of therewrapped input phase angle difference number can be expanded using afunction that is valid over the interval 0<|Δθ_(I)|≦π, and then combinedwith the sign of the input phase angle difference Δθ_(I) to produce theoutput electrical phase angle difference number Δθ_(O). Alternatively,the unwrapped input phase angle difference value can be expanded using arepeating function over the interval −2π≦Δθ_(I) (unwrapped)≦2π. Anexample of such a function is:${\Delta\quad\theta_{O}} = {\pi \cdot ( {\frac{{{\Delta\quad\theta_{I}} - \pi}}{\pi - {\Delta\quad\theta_{I}}} \cdot \frac{{{\Delta\quad\theta_{I}} + \pi}}{{\Delta\quad\theta_{I}} + \pi}} ) \cdot {\sin( \frac{\Delta\quad\theta_{I}}{2} )}}$where Δθ_(I) and Δθ_(O) are the unwrapped signal phase difference valuesmeasured in radians.

Furthermore, the enhancement process can be implemented without a directcalculation of the input signal electrical phase angle difference numberΔθ_(I) which requires the calculation of two arc tangents. In manydigital computation systems, a direct calculation of the arc tangentfunction is relatively computationally intensive, and an enhancementmethod that does not require the arc tangent calculation is desirable.This goal can be accomplished, for example, by using a valueproportional to the tangent of the input signal phase angle differencenumber Δθ_(I) rather than Δθ_(I) itself. Such a value can easily becomputed by using the unit vectors of the input signal vectors A and B.A unit vector is simply a vector that has a magnitude of 1, but the sameangle as that of the original vector. The unit vector can be computed bydividing the complex number representing the input vector by its ownscalar magnitude.

Let A″ and B″ be the unit vectors of A and B. The ratio of the magnitudeof the difference of A″ and B″ to the magnitude of the sum of A″ and B″is equal to the tangent of Δθ_(I)/2. This result can be used directly tocalculate the enhanced output electrical phase angle difference numberΔθ_(O) by use of any enhancement function modified by the replacement ofΔθ_(I) by 2 tan(Δθ_(I)/2) and suitably scaled using methods well knownin the art.

Again referring to FIG. 5, after the input electrical phase differencenumber is enhanced to create an output electrical phase differencenumber Δθ_(O) as shown at 53, the original input electrical phasedifference number is subtracted from the output electrical phasedifference number to create an angle enhancement value, as shown at 54.This value is then divided in two parts and each part is added orsubtracted, as appropriate, to each input signal's phase to thereby movethe signals' phases apart (in the case of expansion) and create a “moreout of phase” condition between the two input signals. The angleenhancement value may be assigned all to one input signal or splitbetween the two input signals in any ratio. One embodiment splits theangle enhancement value equally into two parts at 54, and each half isadded or subtracted, as appropriate, to each input signal's phase,thereby moving the signals' phases apart (in the case of expansion) andcreating a “more out of phase” condition while preserving the sameaverage relative output signal phase. Another embodiment splits theangle enhancement value according to the vector magnitudes so that theresulting output vector's relative phase is identical to that whichwould exist after vector summation if no enhancement had been performed.

To illustrate this aspect of the invention, vector diagram FIG. 7(a)shows that signals A and B are composed of vector sums of the desiredsignal vector component S_(D) and the noise vector components N_(A) andN_(B) respectively. Since the desired signal originates from an on-axis,time aligned source, its component is identical in both signals, asshown by the double vector S_(D). However, since the noise signaloriginates from an off-axis source, the noise components N_(A) and N_(B)are not equal. Although their magnitudes will be equal (barring anydifference in sensor element sensitivity or circuit imbalance), theirelectrical phases will, in general, not be equal, as shown in FIG. 7(a).Thus as shown, the resulting input signal vectors A and B generally willnot be equal in either phase or magnitude.

FIG. 7(b) illustrates the expansion process described above where inputvectors A and B are phase expanded (in the direction of the open arrows)from the input electrical phase difference number Δθ_(O) to the outputelectrical phase difference number Δθ_(O) to become the output vectorsA′ and B′.

After the two input signals are modified, as shown at 56 and 57 in FIG.5, so that their complex number representations have the greater phasedifference, but with their original magnitudes, they are then combinedin the manner of the conventional beamforming method, as shown at 55. Aspreviously mentioned, the two input signals are assumed to result from asensor array system with signal delays as necessary for steering thesystem's sensitivity beam toward the desired signal. Therefore the inputsignals are time-aligned and in-phase for signals arriving from thedesired source, but contain an out-of-phase component for signalsoriginating from off-axis “noise” sources. In keeping with theprinciples of a standard broadside beamforming system, the vectors arethen added without inversion as a vector sum to produce the outputsignal. In this case, the phase-expanded output vectors A′ and B′ arevectorially summed, as shown at 55 in FIG. 5. In other words, each nthbin pair of complex numbers A′ and B′ are vectorially added together toform the complex number placed in the nth bin of the output transform.

This vectorial summation process is illustrated in FIG. 7(c) where theoutput from a conventional beamformer system is shown, compared to theoutput from the inventive system. The signal vector labeled Out is thevector average (the vector sum divided by 2) of the original inputvectors A and B. The major purpose of a noise reduction system is toremove the noise and to put out a signal that is the closestrepresentation of the desired signal. It can be seen by comparison toFIG. 7(a) that the conventional beamformer output vector Out differsfrom the desired signal vector S_(D) both in magnitude and phase. Anyvector difference between the signals Out and S_(D) is a vector (notshown) representing the residual noise left in the output signal afterthe conventional beamformer process is applied.

In contrast, the output vector labeled Out′, which is the vector averageof the signals A′ and B′ that are produced by the inventive method, is avery close match to the desired signal S_(D). The residual noise issignificantly reduced in comparison to that in the output signal of theconventional beamformer, demonstrating the significant noise reductionbenefits of the inventive approach.

Once the data for all frequency bin pairs is processed according to theabove method, a complete output Fourier transform frame is produced. Asshown in FIG. 1 and described earlier, the output Fourier transformframe is then inverse Fourier transformed to produce a processed timedomain output frame. Subsequent processed output frames are thenconcatenated or overlapped-and-added to produce a fully processeddigitized output time domain signal.

Alternatively, the signal information in groups of bins can be firstcombined, for example by vector summation, to produce signal informationon a frequency band basis before the signal processing calculations areperformed. This is often done to reduce calculation costs forapplications where the signal distortions created by band-by-bandprocessing is acceptable. Thus it is contemplated that each calculationis performed on a bin-by-bin or on a band-by-band basis.

FIG. 8 illustrates the beamforming performance of the inventiveapproach. As an example, the performance of a conventional beamformingsystem using two cardioid microphone sensor elements spaced 7-cm apartis shown in FIG. 8(a). It is readily apparent from FIG. 8(a) that thesensitivity beam pattern is essentially that of the cardioid elementsthemselves for low frequencies (below 1000 Hz) where the wavelength islarge relative to the element-to-element spacing s and, thus the arrayaperture is much smaller than one half wavelength. At higher frequenciesthe beam pattern narrows, but as it narrows, side lobes are formed. Forexample, at 3000 Hz, a relatively narrow main lobe is formed, butseveral side lobes are clearly evident. Further, it is obvious that thesensitivity pattern is different for every frequency, and particularlyfor off-axis sounds, the sensitivity is frequency-dependent so thatoff-axis sound signals are changed or “colored”.

In comparison, FIG. 8(b) shows the beam forming performance of a systemin accordance with the invention using the same microphone array, andthe expansion function given by Equation 1 with a sharpness value S_(D)of 10 at 1000 Hz. Not only is the main lobe narrower than that of theconventional beamforming system, but no side lobes are produced.Furthermore, by choosing the sharpness value for each frequency bin tomaintain the shape of the sensitivity pattern the same for allfrequencies, the beam shape for all frequencies is the same, and thereis no “coloration” to the sound from off-axis signals—such sounds areaudibly normal” but attenuated as desired.

The conventional beamforming system can not correct or “flatten” itsfrequency response for off-axis signals for two reasons: 1) there is noparameter available to modify the beam width as a function of frequency(whereas the novel system has the sharpness parameter S), and 2) thebeam patterns show significantly different shapes for each frequency, sothat even if there were a parameter for compensating the beam widthsaccording to frequency, the beam shapes would still not match. In theinventive system, the beam shapes are essentially the same at allfrequencies, allowing for easy frequency compensation by use of a propertapering of the sharpness parameter value vs. frequency, when desired.

FIG. 8(c) shows the beam shape at 1000 Hz for the conventionalbeamforming system in which additional elements have been added to makethe main sensitivity lobe's FWHM (full width at half maximum—a standardmethod of measuring beam widths) equal to that of the novel system underthe same conditions as described for FIG. 8(b). To achieve this equalitycondition, the conventional system needs 13 sensor elements, all spaced7-cm apart for a total aperture (array) size of over 85-cm (assumingreadily available 6-mm diameter electret microphone elements). Thissystem, although large and complex, still does not remove the side lobesof sensitivity.

In comparison, the novel approach provides the 1000 Hz beam patternshown in FIG. 8(d). Not only is the beam pattern free of deleteriousside lobes, but this system requires only two microphone sensor elements(with the concomitant reduction in A/D converters, preamplifiercircuitry and computer processing power) and is less than 9-cm in size.

For greater noise reduction of pickup, the beam can be further narrowedand the sensitivity to off-axis noise sources further reduced. FIG. 8(e)shows the 1000 Hz beam pattern produced by the novel system when thesharpness parameter S is increased to a value of 20. The only limit tothe practical narrowness of the beam is either when the beam is toonarrow to maintain pointing at the desired source, or when theprocessing produces an objectionable level of distortion of the desiredsignal. Practical values for the sharpness parameter in voice-gradecommunication applications using two cardioid microphone elements rangesfrom about 5 to about 50, but is not limited to that range.

In the method of FIG. 5, phase enhancement processing precedes thebeamforming process. Thus, the FIG. 5 method can be readily added to aconventional beamforming system between the sensor electronics and thebeamforming system as shown in FIG. 2(b). As a result, it is clear thatthe above novel phase enhancement approach is highly compatible withconventional beamforming technology for improving the performance ofnearly any beamforming system. For the same reasons, it is also highlycompatible with conventional beam steering and beam tracking systems, asthose of ordinary skill in the art will readily appreciate. Also, justas with conventional beamforming systems, the novel approach is highlycompatible with the use of either omni-directional, bi-directional oruni-directional sensors or sensor arrays. For example, the novelapproach can be used to beneficially combine the outputs of two or moreconventional beamforming array systems. Just as well, two or more of thenovel beamforming systems can provide improved input signals for furthercombination in conventional beamforming systems.

“Wind noise” is an especially troublesome problem in many acoustic voicesignal pickup situations, for example in automobiles for telematicsapplications. Wind noise is different from background acoustic noisesbecause it can not be characterized as coherent sound waves that areimpinging upon the microphone sensors from a distance. Rather wind noiseis characterized by pressure pulses created due to air turbulence in thenear vicinity of, or at, each microphone, and/or microphone port. Thus,it is not possible to determine an angle of arrival for wind noise,since there is no correlation between the electrical phase angles of theindividual sensor signals.

Nevertheless, the inventive apparatus and method disclosed in thisapplication provides significant reduction of wind noise in its outputsignal while preserving the desired voice signal. Because the inputsignal electrical phase angle difference for wind noise can becharacterized as the result of a random process, the electrical angledifference for such noise is statistically uniformly distributed overthe range of possible input signal electrical phase angle differences.Since the inventive process effectively attenuates signals with inputsignal electrical phase angle differences that are away from the apriori known difference (typically 0 degrees) for the desired signal,wind noise is also effectively attenuated over much of the input signalelectrical phase angle difference range.

Such operation is highly desirable in acoustic sensor systems where windor moving air is a problem due to the “wind noise” it creates.

A problem encountered with conventional beamforming technology is theneed for sensitivity matching of the sensor signals to achieve maximumnoise reduction performance. Although adequate matching of the sensoramplifiers and A/D channels is relatively straightforward, matching ofthe sensors themselves is not. To continue the use of the acoustic audioexample, microphone elements are difficult and expensive to match, andmaintaining the match across temperature changes and aging is evenharder. Further, matching of sensor sensitivities at one frequency ispossible, but matching over all frequencies is very difficult, evenwithout taking temperature fluctuations and aging effects into account.

Some beamforming systems attempt to automatically match the sensorsignals by applying Automatic Gain Control (AGC) amplification for eachsensor channel, controlled by factory measured sensitivity differencessaved in a memory system to be applied as a correction later duringoperation or by actively and periodically injecting matched energysignals into the sensors and correcting any sensitivity differencesbased upon measurements of the result of these “probe” signals.

As shown in FIG. 2(c), the signal sensitivity matching methods describedabove are usually applied to the sensor signals 22 before those signalsare processed. Thus, when used with the novel system of the presentinvention, the sensitivity matching circuit 26 is located to precede thephase enhancement circuit 24, as shown. Alternatively, signal matchingcan also be applied after phase enhancement, since the enhancementcircuit 24 only modifies the phase of the input signals and does notalter their magnitudes. Further, the equalization circuit 26 need notonly be used for magnitude (amplitude) matching, but in addition canprovide frequency equalization, when required.

Each of the prior art sensitivity matching methods has drawbacks. TheAGC method can correct sensitivity differences at one frequency, but cannot match the sensitivity over all frequencies. It also takes time toadjust, and this correction delay can be a problem in systems requiringrapid responses to incoming signals. The factory measured matchingmethod can operate over frequency and without delay, but can not tracksensitivity changes due to temperature, humidity or aging. The probesignal method requires that the beam forming system be taken off-lineduring the periodic signal injection phase of operation. In addition,all of these methods add significant cost and complexity to the system.

To demonstrate the need for sensitivity matching, consider the use of aconventional beamforming system to detect speech in noise. Speechconsists of short bursts of voice sounds separated by periods ofquiet—the pauses between speech bursts. It is critical that abeamforming noise reduction system reduce the effects of off-axis noisesduring the speech pauses, since at those times there is no voice signalto mask the effects of small amounts of residual noise and any residualnoise becomes quite audible.

Referring again to FIG. 7(a), during a speech pause, the desired signalS_(D) becomes zero and the input signals A and B consist only of thenoise vectors N_(A) and N_(B), as shown in FIG. 7(d). In this case, thesignal is only noise and the desired result is an output of zero.

When the input signals are combined in a conventional beamforming systemwith matched sensor signals, the resulting output signal is reduced asexpected, although not to the desired value of zero. This is shown inFIG. 7(e) by the average output vector labeled Out which is the resultof conventionally beamforming just the noise signals A=N_(A) andB=N_(B).

However, with a sensor signal mismatch, the residual noise in thesystem's output signal is significantly increased. Typical microphoneelements are available with sensitivity matching at 1 kHz of ±3 to ±4dB. Thus, in the two sensor case, if one sensor is at the low end of thesensitivity distribution and the other at the high end, the two sensorsignals can be mismatched by a sensitivity difference of 2:1 or more.FIG. 7(f) shows the output vector that results from a conventionalbeamforming system where the sensors are mismatched such that sensorsignal A is reduced by 3 dB and the sensor signal B is increased by 3 dBas compared to those shown in FIG. 7(e). In this case, the conventionalbeamformer output signal vector Out is significantly increased inmagnitude and is significantly altered in phase. This effect results inincreased audibility of the off-axis noise put out by the conventionalbeamforming system.

FIG. 7(e) also shows the residual output noise after processing by thenovel system of this application, again assuming matched signalmagnitudes. As is shown in FIG. 7(d), the relatively large input signalphase angle difference number Δθ_(I) means that the expanded outputphase difference number Δθ_(O) will be very close to 180 electricaldegrees. Thus, the output signal vectors A′ and B′ will be essentiallyout-of-phase but of the same magnitude, as shown in FIG. 7(e). When thiscondition is achieved, the two signals A′ and B′ will cancel each otherwhen vectorially summed at 55 in FIG. 5, resulting an essentially zerolength output vector as shown in FIG. 7(e) by the dot labeled Out′.Thus, when the sensor signals are well matched in sensitivity, the novelinvention achieves the desired result of very low output for suchnoise-only signals. Compared to the residual noise output vector Outprovided by the conventional beamforming system, the residual noiseoutput vector Out′provided by the novel system is much smaller—that is,the residual noise in the output of the novel system is more greatlyreduced.

It is to be appreciated that the novel beamforming system utilizes phaseenhancement functions that include a sharpness parameter S that allowscontrol over the beam width. Therefore, the value of the sharpnessparameter can be chosen or controlled so as to produce beneficial newcharacteristics for a beamformer. For example, adjusting the sharpnessparameter in response to an increase in the noise level can be used tonarrow the beamwidth as more noise rejection is needed. The value of Scan be automatically adjusted by detecting the noise in the outputsignal and adjusting the value to maintain, for example, a specifiedoutput signal-to-noise ratio.

Alternatively, in applications where the noise is known to have certainfrequency characteristics, for example where most noise consists of lowfrequencies, the value of S can be adjusted to produce a wide beam atthose frequencies in order to maintain the best signal quality, while anarrow beam can be created at other frequencies to maximize rejection ofthose noises. Such frequency tapering of the beamwidth can be fixed,manually adjusted or made adaptive or automatic by controlling the valueof the sharpness parameter S in response to the variation of a controlsignal. There are many such ways to apply the extra freedom allowed bythe sharpness parameter, and all are contemplated to be in accordancewith the invention.

Further in accordance with the invention, a novel algorithmic matchingmethod is provided to avoid the drawbacks of slow response, changetracking, added cost and complexity associated with the conventionalbeamformer system signal matching methods. This novel matching systemprovides the benefits of an instantaneous sensitivity match for allfrequencies and over temperature, humidity and sensor aging conditions.Furthermore, this novel signal matching process can be applied tovirtually any array system where matched signal sensitivities are neededor desired, and it is not limited to use with the novel beamformingsystem provided herein, although it works well in this system to assurematched signals for maximum noise reduction performance.

FIG. 9 shows the inventive signal sensitivity matching system asimplemented within the framework of the novel beamforming system. Againdescribed in the context of a two-sensor array although no suchlimitation is intended, input signals A and B are first separated intotheir phase and magnitude components at 91 a and 91 b. Circuit blocks92-97 correspond to the same blocks labeled 52-57 in FIG. 5, andrepresent substantially the same process steps. The signal amplitudematching is created by new circuit block 98 wherein the two input scalarsignal magnitudes |A| and |B| are combined to create a new common scalarmagnitude value GM equal to the mathematical mean of |A| and |B|. Inthis example, the geometric mean value is used. This new scalarmagnitude value is then used at 96 and 97, combined with the expandedphase values Δθ_(AO) and Δθ_(BO) to produce the phase expanded outputsignals A′ and B′.

This inventive method of compensating for sensor sensitivity mismatchesand sensor signal path differences uses the process of reassigning tothe magnitudes of the expanded electrical phase angle vectors themathematical mean value of the individual magnitudes of the inputvectors. There are numerous types of mathematical means, for example thearithmetic, root-mean-square (ms), geometric, harmonic and others. Forthe purpose of this invention all mathematical means are applicable, anda particular mathematical mean may be used as the design requires.

Use of the arithmetic mean, defined as$\frac{1}{N}\quad{\sum\limits_{i = 1}^{N}{S_{i}}}$(where S_(i) is the signal from the ith sensor and N is the total numberof sensors) produces little attenuation for highly mismatched signals,and does not extinguish the output signal if a sensor fails completely.The rms is even more forgiving in its capability to prevent attenuationof highly mismatched signals and also does not extinguish the output forsensor failures. The fail-safe characteristic of these mathematicalmeans makes them very desirable for many real-world applications where areliable system must continue to operate, albeit with reduced efficacy,in the event of sensor failure.

However, highly mismatched signal amplitudes are also created byundesirable multi-path, clutter or reverberation artifacts andadditional attenuation of such signals is desirable in such situations.Use of the harmonic mean, defined as${N \cdot ( {\sum\limits_{i = 1}^{N}\frac{1}{S_{i}}} )^{- 1}},$creates relatively aggressive attenuation of such undesirable artifacts.This artifact reduction capability makes the harmonic mean a good choicefor applications where clutter is a significant problem.

In contrast, the geometric mean, defined as$\sqrt[N]{{{S_{1}} \cdot {S_{2}}}\quad{\ldots \cdot {S_{N}}}},$provides a good compromise between the attenuation of such undesirableartifact noise signals while preserving the quality of the desiredon-axis signal. In the case of human signal perception, such as in sight(light) or hearing (sounds and speech), a logarithmic mean is desirable,and the geometric mean provides this characteristic. For example, if onesensor has +X dB (greater) sensitivity than nominal while the other has−X dB (lower) sensitivity than nominal, then use of the geometric meanwill provide an output magnitude equal to that provided by a matchedpair of nominal sensitivity sensor elements, so that mismatches becometransparent to the user.

Although the system designer will choose the preferable mean for theparticular application being addressed, for acoustic voice signals thegeometric mean is to be preferred.

A valuable element in this new signal sensitivity matching system is theuse of a mathematical mean magnitude value to replace the individualmagnitudes of the input signals. If applied to a conventionalbeamforming system, the phase enhancement process would be bypassed, andthe original input signal phases, in this case θ_(A) and θ_(B), would beused at 96 and 97 instead.

Referring again to FIG. 2(c), this novel signal sensitivity matchingmeans can be applied before, or after, the phase enhancement process. Inthis figure, circuit block 26 is shown before the phase enhancementblock 24, but the locations can be reversed without affecting theperformance of the overall system. Indeed, if the phase enhancementcircuit block 24 is eliminated, then it is easy to see that the novelsensitivity matching process disclosed above can be easily added to aconventional beamforming system between the sensor electronics and thebeamforming system.

Benefits of this novel sensitivity matching system are its continuousmatching capability, virtually instantaneous matching, ability tocontinuously correct at all frequencies in real-time, without delay ordead-time, to eliminate the effects of mismatch, drift, aging,temperature, humidity and all other causes of sensitivity changes.Applicability includes radio, sonar, audio, radar, medical imaging,optics, and other array systems where matched sensors are required.

As shown in the vector diagram of FIG. 10, when the signal magnitudesare matched, the input signal vectors A and B form an isoscelestriangle. In the conventional beamforming system, the output signal Outis created by calculating the vector average of A and B, and theresulting output signal vector bisects the triangle, as shown. Thus, aright triangle O-B-Out is formed, where the magnitude of the outputsignal vector Out is given by: $\begin{matrix}{{{Out}} = {{{A} \cdot {\cos( \frac{\Delta\quad\theta_{I}}{2} )}} = {{B} \cdot {{\cos( \frac{\Delta\quad\theta_{I}}{2} )}.}}}} & (2)\end{matrix}$

Similarly, in the novel beamforming system, when the signal magnitudesare matched, the input signal vectors A′ and B′ form another isoscelestriangle. The output signal Out′ is created by calculating the vectoraverage of A′ and B′ (at 55 in FIG. 5, or 95 in FIG. 9), and the newoutput signal vector Out′ bisects this triangle. Thus, a right triangleO-B′-Out′ is formed, where the magnitude of the output signal vectorOut′ is given by: $\begin{matrix}{{{Out}^{\prime}} = {{{A^{\prime}} \cdot {\cos( \frac{\Delta\quad\theta_{O}}{2} )}} = {{B^{\prime}} \cdot {{\cos( \frac{\Delta\quad\theta_{O}}{2} )}.}}}} & (3)\end{matrix}$

When phase-expansion is applied to the input signal electrical phaseangle difference, the magnitude of this output vector Out′ is alwaysless than that of the conventional beamformer output vector Out,although the phase of the output signal is unchanged. Thus, with matchedsignal levels, the phase expansion process of the novel noise reductionbeamforming system reduces the magnitude, but retains the phase, of theoutput signal produced by the conventional beamforming system. Thisreduction in magnitude is shown in FIG. 10 as the difference in vectorlengths 101.

A more computationally efficient approach utilizes this signalattenuation characteristic directly, instead of first calculating theexpanded phase vectors A′ and B′. FIG. 11 illustrates this approach. Asshown in FIG. 11, the input signals 112 from the sensor array 111 areamplitude matched at 116 if not inherently matched. Matching can becreated by use of conventional array matching methods or by use of thenovel mathematical mean matching method described above. The matchedsignals are then vector summed in a conventional beamformer 113 beforebeing attenuated at 118 by an attenuation amount provided from circuitblock 117. The attenuation amount is determined from the measured inputsignal electrical phase angle difference number Δθ_(I) at circuit block117, as will be described. The attenuation amount is not dependent uponthe magnitudes of the input vectors, or upon their absolute phases, butonly upon the input signal electrical phase angle difference value ornumber.

Since the output electrical phase angle difference number Δθ_(O) isdirectly calculated from the input electrical phase angle differencenumber Δθ_(I) (for example, as described by Equation 1), a morecomputationally efficient method for creating the output signal Out′ isto compute the attenuation as though the input signals are matched, andthen to apply this attenuation to the output of the conventionalbeamforming system. Although, without signal matching, or with theconventional signal matching methods, the input signal magnitudes mightnot be well matched, this computationally efficient method can still beapplied, but may result in an error in the phase of the output signal.

Recognizing that, for audio applications, the human ear does not readilydistinguish the phase of signals, this inconsequential phase errorbecomes unimportant. Thus, for an audio communication device, the phaseof the output signal can be slightly altered without impairing theefficacy of the system's noise reduction. Indeed, the slight deviationin output phase used in this method is likely not an issue with mostcontemplated applications, such as for example sonar, radar, optical,radio antenna systems and the like. However, with the novel signalmagnitude matching method, phase error is not a problem, since theoutput signal phase will be perfectly retained.

As seen in FIG. 10, the amount of attenuation to be applied is the ratioof the magnitudes of the output vectors Out′ and Out. Since the signalsA′ and B′ are assumed to be sensitivity matched and equally expanded,the output vectors Out′ and Out have the same electrical phase angle, asshown in FIG. 10. Therefore, the ratio of their magnitudes, fromEquations 2 and 3, becomes a simple scalar attenuation value defined as$\begin{matrix}{{Attn} = {\frac{\cos( \frac{\Delta\quad\theta_{O}}{2} )}{\cos( \frac{\Delta\quad\theta_{I}}{2} )}.}} & (4)\end{matrix}$Since Δθ_(O) is a function of Δθ_(I), the attenuation value is only afunction of Δθ_(I).

FIG. 12 shows a flow diagram of how this attenuation ratio can be usedto provide another way of implementing the inventive noise reductionmethod. Using Equation 4, plus a relevant phase enhancement function fordetermining the expanded output electrical phase difference numberΔθ_(O), an attenuation value can be determined from the input electricalphase angle difference number Δθ_(I). This attenuation value is thenused for modifying the output from a conventional beamformer to producethe same noise reduced output as is produced by the phase enhancementmethod. In this method, the two input signals are first vector summed at125 to produce an un-attenuated intermediate signal. Using only theinput signal electrical phases at 121 a and 121 b, the input electricalphase difference number Δθ_(I) is calculated at 122 and subsequentlyused, along with a phase enhancement function or lookup table, tocalculate the output electrical phase difference number Δθ_(O). Theattenuation value is then calculated according to Equation 4 at 128.

When the input signals are from the desired on-axis source, the twoelectrical phase difference numbers are equal and the attenuation valueequals unity, resulting in this desired signal being passed withoutattenuation. When the input signals are from an off-axis noise source,the two electrical phase difference numbers are unequal, with the outputelectrical phase difference number always being larger than the inputelectrical phase difference number. Because the phase difference numbersare divided in half, they lie on the interval −π/2≦Δθ<π/2 and alwayshave the same sign. Thus, the attenuation value for such signals,according to Equation 4, will be less than one, and will decrease towardzero as the input noise signal arrival azimuth angle increases away fromthe array axis toward 90 degrees.

As an example of this attenuation characteristic, FIG. 13 shows a graphof the attenuation value created, using Equation 4, and the phaseenhancement function of Equation 1 (which is graphed in FIG. 6(a), fordifferent values of the sharpness parameter S). The horizontal axis ofthis graph is the input electrical phase difference number Δθ_(I), whilethe vertical axis is the attenuation value. Curves 130, 131 and 132illustrate the attenuation value as a function of the input signalelectrical phase difference number for sharpness values of 5, 10, and 20respectively. Notice that the attenuation value is equal to unity at aninput electrical phase difference number of zero degrees, since thisrepresents the phase difference for a desired signal. As the inputelectrical phase difference increases away from zero in eitherdirection, the attenuation increases—that is, the attenuation valuedecreases toward zero.

Again referring to FIG. 12, after calculation of the attenuation valueat 128, the conventional beamformer intermediate signal vector from 125is multiplied by the scalar attenuation value at 129 to produce thefinal attenuated output signal. Thus, as the input electrical phasedifference number increases away from zero in either direction, theconventional beamformer output signal is attenuated, since such inputsignals must be from off-axis noise sources. As with the phase expansionprocess of the inventive method, this attenuation process similarlyremoves the effects of off-axis noise sources in the output signal ofthe system.

It should be noted that this attenuation method saves computation at 121a and 121 b by not requiring the calculation of the magnitudes of theinput signals A and B, and further saves the calculation of the phaseexpanded output vectors A′ and B′. However, it still requires thecalculation of the expanded output phase difference number Δθ_(O). Afurther saving in computation can be made by using an attenuationfunction, rather than a phase expansion function such as thosepreviously described.

Although less intuitive, this very computationally efficient approachaccomplishes the same noise reduction as the previously describedapproaches of the inventive system. This approach will be described withreference to FIG. 14.

Remembering that the input signals A and B are Fourier transformbin/band values, the inputs are vectors represented by complex numberswith a real and an imaginary part. At circuit 140, the square root ofthe ratio of the magnitude of the input signal A to the magnitude of theinput signal B is calculated—that is, the output of circuit 140 is thescalar value: ${{Output}\quad 140} = \frac{\sqrt{A}}{\sqrt{B}}$

This scalar value Output 140 is used at circuit 141, to divide the inputvector A, whose magnitude is |A|. The result is that the output vectorsignal from circuit 141, the vector signal A′″, has a magnitude equal tothe geometric mean of the magnitudes of the two input vectors A and B,but the electrical phase angle of input vector A. The scalar value from140 is also used at circuit 142 to multiply input vector B, resulting inthe vector signal B′″, whose magnitude is also the geometric mean of themagnitudes of the two input vectors, but whose electrical phase angle isthe same as that of input vector B. It will be appreciated that themethod shown in FIG. 14 inherently provides geometric mean magnitudeequalization which serves to correct for the unmatched character of thetwo sensors.

The two geometric-mean magnitude matched vector signals A′″ and B′″ arethen summed at 144 to obtain the un-attenuated intermediate outputvector, while the vector difference of A′″ and B′″ is obtained at theoutput of circuit 143. Remembering that the vector difference divided bythe vector sum is equal to the imaginary operator (√−1) times thetangent of half the angular difference, the circuit 145 computes thisratio—that is, the signal T is:$T = {\frac{A^{''\prime} - B^{''\prime}}{A^{''\prime} + B^{''\prime}} = {{\mathbb{i}} \cdot {{\tan( \frac{\Delta\quad\theta_{I}}{2} )}.}}}$The imaginary operator is removed at circuit 146 by taking the magnitudeof T, resulting in a scalar value equal to the tangent. This tangent ofone half of the input electrical phase difference number Δθ_(I) is thenused at circuit 147 to calculate the attenuation by application of anattenuation function or lookup table. Once the attenuation value isdetermined from the function or table, it is applied to the intermediateoutput vector signal from circuit 144 by multiplying the vector signalby the attenuation value. This produces the final output from the noisereduction process.

FIG. 15 shows a graph of, and the defining equations for, some typicalattenuation functions that can be used with this novel beamforming noisereduction system. As was discussed with reference to various curvesshown in FIG. 13, the attenuation value for an input electrical phasedifference number of zero degrees is one—in other words such a signal ispassed without attenuation. Thus, a desired signal, one that originatesfrom a position along the array sensitivity axis, is not attenuated.However, as the input electrical phase difference number increases ineither direction away from the zero value, additional attenuation iscreated, since the attenuation value drops below the value of one andtoward the value of zero. Thus, signals originating from off-axis noisesources are attenuated.

As noted by curve 151 of FIG. 15, the attenuation value need not reachzero for any input electrical phase difference, including an inputelectrical phase difference of 180 degrees. Furthermore, the attenuationneed not fall monotonically to either side of zero degrees. Indeed manyother functions and curves can be used, as long as there is attenuationfor at least some input electrical phase difference numbers away fromzero.

The attenuation function shown by the solid graph curve 150 of FIG. 15,is especially interesting for use with the method described with respectto FIG. 14. This is because, as shown in the defining equation for thatcurve, the attenuation value is determined by the tangent of one half ofthe input electrical phase difference number, Δθ_(I), which is thesignal that is inherently available at the input of circuit 147 in FIG.14.

This function also promotes efficient calculation because the sharpnessparameter is used in a simple multiplicative manner, rather than as apower. When this, or a similar, attenuation function is used in themethod of FIG. 14, very simple and efficient computation results.Functions incorporating multiplicative use of the sharpness parameterare highly desirable because of their low computational powerrequirements.

The forgoing discussion has described a method of determining applicableattenuation values by calculation of the attenuation values fromfunctions. Alternatively, the attenuation values can be obtained from alookup table of pre-calculated values at circuit block 147. In such animplementation, the computational overhead of calculating the valuesfrom a function is eliminated. This method provides even greaterreal-time computational efficiency, although at the expense of reducingthe ability to make real-time changes to the attenuation table values inresponse to changing conditions.

This discussion has addressed only a few examples of the possibleattenuation formulas and curves, and is not intended to be limiting.Formulae that include the point 0,1, and curves that pass through thepoint 0,1, and increase the attenuation at other points conform to oneaspect of the invention. Formulae and curves that maintain noattenuation at some other selected point, and increase the attenuationat other points conform to another aspect of the invention. Inaccordance with a further aspect of the invention, attenuation is onlyapplied for some input phase angle difference number values Δθ_(I). Inpractice, attenuation will likely be applied, to a greater or lesserextent, to most values, although it will be recognized that there is norequirement that attenuation be applied to most or even a substantialportion of the values. Further, for systems in which a symmetricalattenuation function is applicable, the calculation of the attenuationcan be simplified or the look-up table minimized by using just themagnitude of the input phase angle difference number values Δθ_(I).Further, re-wrapping of the input phase angle difference value Δθ_(I) isunnecessary when attenuation functions that repeat over intervals of πto 2 π and −2 π to −π are used. The above are exemplary modes ofcarrying out the attenuation method and are not intended to be limiting.It will be apparent to those of ordinary skill in the art thatmodifications thereto can be made without departure from the spirit andscope of the invention

It should be noted that the maximum value of the attenuation value neednot be equal to one. If the maximum value is made equal to a lower valueso that the attenuation values are scaled to fit the range from zero tothis lower value, then the beam shape enhancement will be retained, butthe overall sensitivity will be lowered. Similarly if the maximum valueis made equal to a higher value than one so that the attenuation valuesare scaled to fit the range from zero to this higher value, thenalthough the beam shape is again unchanged, the overall sensitivity willbe increased. It should be noted that the term “attenuation” is stillapplied even when the attenuation value is greater than one, although insuch a situation the opposite of attenuation is taking place. In otherwords, a signal that is multiplied by an “attenuation” value greaterthan one is actually being magnified (undergoing a gain), rather thanattenuated. Nevertheless, the term attenuation will be used herein. Thisleads to a simple method for gain control that can be easily integratedinto the attenuation method of implementing the novel beamformingprocess. Such gain control, for example, can be used for Automatic GainControl of such a system with an appropriate control signal, as is wellknown in the art. There are many applications where gain control,including AGC, is of great benefit.

Further in accordance with the invention, a phase difference other than0° can be used as the phase difference for which no phase enhancement orattenuation occurs, using a function that provides phase expansion orattenuation elsewhere. In this manner, the direction of maximumsensitivity is steered to an angle corresponding to an azimuthal angleother than that which produces 0° of electrical phase difference. Withother parameters held constant, changing the input signal electricalphase difference at which no phase enhancement or attenuation is appliedshifts the azimuthal angle of maximum sensitivity of the sensor system.

The attenuation curves 130, 131 and 132 of FIG. 13 demonstrate noattenuation at the on-axis “look” direction of 0 degrees for the inputsignal electrical phase angle difference number, but demonstrate signalattenuation for input signal electrical phase angle difference numbersthat are away from 0 degrees. Another use for the inventive apparatusand method is to provide a new way of beamsteering. Conventional methodsof beamsteering require the application of time-delay techniques, and/ortheir equivalent in the frequency domain. Instead for example, if thecurves shown in FIG. 13 are displaced laterally so that zero attenuationoccurs at an angle other than 0 degrees, as illustrated by curve 133,then the effective beam is moved, or “steered”, to this new angle wherethe attenuation is zero.

Such steering can be accomplished in a fixed manner, or dynamically inreal time by applying an attenuation function that shifts its peaklaterally in real time as needed, or in response to the control signalfrom a beam tracking system (not shown). Remembering that Equation 4shows that there is a corresponding phase expansion function for everyattenuation function and vice versa, it will be obvious to one skilledin the art that this new form of beamsteering can also be accomplishedby utilization of an appropriate (or corresponding) phase enhancementfunction along with the phase enhancement methods described above.

Further, there are numerous beamformer applications in which multiplesimultaneous beams are required, for example in sonar and radarapplications. Using attenuation functions with two or moreno-attenuation peaks, but with finite attenuation between those peaks,can produce multiple-beam pattern sensitivity peaks or beams. Similarly,those beams can be steered, and each can be steered independently, bydynamically moving the lateral locations of the attenuation functionpeaks as required, for example, in response to an appropriate controlsignal (which could be a beamtracking control signal). Also as describedabove, this multiple beam apparatus and method can be accomplished withthe phase expansion method previously described in detail by use of thecorresponding phase expansion function.

An example of an excellent application for this technique is thesuper-resolution beamformer, where the deleterious effect of the signalsensitivity side lobes in a first beamformer's sensitivity pattern isreduced or cancelled by adding an appropriately scaled and invertedsignal from a second beamformer of the type just described to the signalfrom the first beamformer. To accomplish this cancellation, the secondbeamformer's sensitivity pattern would mimic the side lobes of thefirst. Thus, the side lobes of the first beamforming system areeffectively cancelled using this means, leaving just the narrow mainlobe of sensitivity.

All beam forming systems create some amount of distortion of the desiredsignal. As such a system becomes more aggressive—that is, as it producesa narrower sensitivity beam pattern—the distortion increases. For theinventive system, the distortion that is created becomes measurable, butonly for high values of the sharpness parameter, S. Thus, it is valuableto attempt to minimize the value of the sharpness parameter, S, wheneverpossible, in order to minimize the distortion, and the tradeoff ofincreased distortion can be balanced with increased sharpness parameterin accordance with the particular application.

As described above for implementing the inventive signal process, boththe phase expansion and the phase-based attenuation methods have beenshown with symmetrical improvement functions, for example as shown inFIGS. 6, 13 and 15. However, the improvement functions, whetherimplemented in direct calculation form or in look-up table form, neednot be symmetrical. Certain applications can benefit from the use of anasymmetrical beam pattern; for example in an optical application whentrying to resolve the signal from a faint star next to a bright star. Anarrower beam or greater attenuation to the side of the brightinterfering star can attenuate the interference from this “noise”source, while providing the normal beam or attenuation in all otherdirections, thus minimizing the distorting effect of the high values ofsharpness used to produce the narrower beam. In this manner enhancementor attenuation is conducted asymmetrically about a selected phase angledifference, which, in the case of curves 130-132 (FIG. 13) for example,is zero degrees, while in the case of curve 133, it is a value otherthan zero degrees.

Such an asymmetrical directionality improvement can be created by, forexample, using one sharpness value for positive input signal phasedifference values while a different sharpness value is used for negativeinput signal phase difference values. Similarly, one improvementfunction or table can be used for the positive side while another isused for the negative side.

Further, the value of the sharpness parameter S can vary with frequency.For example, using a single value for all frequencies creates a beampattern that is relatively broad at low frequencies, but becomesnarrower at higher frequencies. This is because the wavelength of asignal varies inversely with frequency, and therefore the inputelectrical phase difference for an off-axis signal varies linearly withfrequency.

The beam width can be made equal for all frequencies by correcting forthis effect. One means of correcting for this effect is selecting adifferent sharpness parameter value for each frequency in such a manneras to compensate for the change. For example, the uniform beam-widthshown in FIG. 8(b) occurs when the sharpness parameter, S, is adjustedas the inverse of the frequency difference. When the 1 kHz sharpnessparameter is set to a value of 10, at 500 Hz the same beam width iscreated by a sharpness value of 20, and at a frequency of 2 kHz therequired sharpness value is 5. Therefore, by choosing the sharpnessparameter value as a function of frequency, virtually any desiredfrequency response for off-axis sensitivity can be created.

For applications in which a symmetrical improvement applies or isdesirable, the computational cost can be reduced by exploiting thesymmetry. Using the magnitude of the input signal phase difference valueto determine the amount of noise improvement can remove the need forcalculating the signum sgn(Δθ_(I)) function, for example as in Equation1, or can reduce the size of a look-up table by a factor of two.

Generally, the microphone spacing (s, in FIG. 4) would be one-halfwavelength or less at the highest frequency of interest. This is becausethe calculated input signal electrical phase difference should notexceed ±180°. When the difference exceeds ±180°, the value becomesambiguous. For example, if the sensor spacing were equal to a fullwavelength and a noise source was located at an azimuth angle of 90°,the true value of the input signal electrical phase difference, Δθ_(I),would be 360°. However, the calculation of the input signal electricalphase difference would create a mathematical value of 0° and theresulting signal then would not be attenuated. The result is that theresulting sensitivity beam has side lobes at frequencies exceeding thatat which the spacing is one-half wavelength. This is not necessarilyundesirable in some applications, for example, in applications where allimportant noise sources have frequency content below the frequency wherethe sensor spacing is one-half wavelength, but the desired source hascontent above that frequency.

However, for other applications, without a means of calculating the trueinput signal electrical phase difference, such larger sensor spacingscan be a problem. FIG. 16 shows a method for both extending the novelmethod to linear broadside arrays of greater than two elements, but alsoa means for resolving the input signal electrical phase differenceambiguity created by greater sensor spacings. FIG. 16 shows an array,160, having three sensor elements, A (162), B (164) and C (166), whereinthe sensor-to-sensor element spacing, s, is one-half wavelength, but thearray width is one full wavelength. Here the system determines the inputsignal electrical phase differences between all sensor signal pairs,A-B, B-C and A-C, wherein the inner pair electrical phase differences,A-B and B-C, are always between ±180°, but the outer pair differenceranges over the interval ±360°. The inner pair electrical phasedifference values can be averaged or used singularly as a coarse measureof the azimuthal angle of arrival, while the outer pair electrical phasedifference value is used as a fine measure of the angle of arrival.While the inner pair phase difference value(s) resolves the ambiguity,the outer pair phase difference value is used to produce the phaseexpansion or phase-based attenuation of the noise. Thus, a greaternarrowing of the effective beam 168 can be achieved without additionaldistortion of the desired signal. This method can be extended to anysize array with any number of elements, whether spaced uniformly ornon-uniformly.

In the aforementioned configurations, the novel technology is equallysensitive to all signal sources lying on the sensitivity axis Iregardless of their distance from the array, and only attenuates signalsbased on their angle of arrival. However, in many applications it isdesirable to also provide a means for accepting only signals originatingfrom a specific distance, or “range”. FIG. 17 shows two methods forproducing a range sensitive beam pattern in accordance with theinvention.

In FIGS. 17, 18 and 19, circuit blocks labeled PROCESS implement thenovel beamforming process of this invention, using any of the disclosedmethods including the phase enhancement and/or attenuation methods.Similarly, circuit blocks labeled Δθ ENH implement just the phaseenhancement process of this invention using any of the disclosedmethods. Such uses of the inventive processes are not intended to belimiting.

In FIG. 17(a), a desired source, S_(D), is shown located within region175 at a distance from the array formed by the four sensors A, B, C, andD. Sensors A and B, along with PROCESS 171, form a first one of theinventive beam forming systems that produces beam 172. Sensors C and D,along with PROCESS 173 form a second one of the inventive beam formingsystems that produces beam 174. The sensors are all located at the samedistance from the desired source S_(D) (as shown) or their signals maybe time aligned for the desired source, using conventional signal timealignment techniques. The signals put out from these first and secondbeam forming systems are combined in a third one of the inventiveprocesses, PROCESS 177, to produce the final output signal. In such away, only signals originating from sources within the sensitive region175 are detected, while the signals from “noise” sources located outsidethe sensitive region 175 are attenuated. Therefore, both angle and rangeresolution are achieved by such a system.

In such a system, the circuit blocks labeled PROCESS need not be thesame. For example, the process at 171 and 173 could implement the phaseenhancement method, while the process at 177 could implement theattenuation method. Further, one or more of the process circuit blockscan be implemented as a conventional beamformer.

FIG. 17(b) shows a simpler method for creating range resolution usingthe inventive method. Similar portions of FIG. 17(b) are labeled withthe same designations as used in FIG. 17(a). Here the sensors lie on astraight line, and time delay circuits 178 and 179 are used to steer thetwo beams 172 and 174 inward, as shown. Thus, all sensor signals becometime aligned by this means. Alternatively, the sensors can be located atequal distances from the desired source, as shown in FIG. 17(a) therebyremoving the requirement for the time delays shown in FIG. 17(b). Whenthe signals produced by the sensors are thus time aligned, they can beused in a single beamforming process PROCESS 177 where the input signalelectrical phase difference values for signal pairs A-B and C-D arefirst determined. In addition, all four signals arriving at the PROCESS177 are vectorially added together as in a conventional beam formingsystem to form an intermediate output signal. The largest of theelectrical phase difference values is then used to determine theattenuation to be applied to the intermediate output signal in a mannersimilar to that described in reference to FIGS. 12 or 14. Once theattenuation is applied, the result is the final output signal shown inFIG. 17(b). Alternatively the phase expansion techniques, as describedin reference to FIGS. 5 and 9 can be first applied separately to signalpairs A-B and C-D, before the four resulting phase expanded signals arethen vectorially added together to produce the final output. In thismanner, a range sensitive system can be realized in accordance with theinvention.

As so far described with respect to linear broadside arrays, theinventive method produces an effective reduction in the width of asensor beam produced by an array of sensors. FIG. 18 shows threedifferent means of using the novel technology for creating a “pencil”beam—that is, a beam with both reduced azimuthal (width) and reducedelevational (height) extent. Although three different arrangements areshown, they are intended only as examples of the invention, and are notintended to be limiting.

FIG. 18(a) shows a four sensor method in which the signals from sensorsA and B are used by a first one of the inventive processes PROCESS 181to produce a first intermediate signal 182 representing a firsteffective sensitivity beam that is narrow in the X direction, butrelatively wide in the Y direction. Simultaneously, the signals fromsensors C and D are used by a second one of the inventive processesPROCESS 183 to produce a second intermediate signal 184 representing asecond effective sensitivity beam that is narrow in the X direction, butrelatively wide in the Y direction. The phase difference between the twointermediate signals 182 and 184 contains information about theelevational angle of arrival for signals that are off-axis in the Ydirection. A third one of the inventive processes PROCESS 183 uses thiselevational angle of arrival information contained in these twointermediate signals to produce a final output signal representing afinal pencil-shaped sensitivity beam that is narrow in both the X andthe Y directions.

Although relatively simple to create and understand, the pencil beammethod of FIG. 18(a) is complex, uses a relatively high number ofcomponents and requires relatively high computational power. To reducethis cost, another pencil beam method is shown in FIG. 18(b). Here athree sensor array consisting of sensors A, B and C in a triangularconfiguration is shown. Preferably, the sensor elements are arranged inan equilateral triangle configuration, but that configuration is not alimitation for the purposes of this invention and other three sensorconfigurations are contemplated. The three sensor signals are used bythe inventive process shown at 186. Although the phase expansion methodof implementing the novel system can be used at 186, one of theattenuation processes, such as those described in connection with FIGS.12 and 14, will be described.

First the process calculates the absolute value of the input signalelectrical phase difference values for sensor signal pairs A-B, B-C andC-A. Then either the mean value of these three input signal electricalphase difference values is selected or the largest value is selected,and the resulting input signal electrical phase difference selection isused to determine the attenuation amount to be applied to the vectorialaverage of the three sensor signals. This attenuated vectorial averageis the final output signal for the system, and it represents a beampattern that is narrow in both the X and Y directions, as desired.Although any mathematical mean can be used, in general the average valuewill be desirable. This pencil beam system is significantly simpler andless costly than the four sensor system described with respect to FIG.18(a). However, there is an even simpler system configuration forproducing a pencil beam using the inventive technology.

FIG. 18(c) shows such a system. A two-sensor array is formed by sensorelements A and B configured as an end-fire array along the sensitivityaxis L By time delaying the sensor signal from the front sensor A usingtime delay circuit 187, the delayed signal 188 from sensor A and thedirect signal from sensor B arrive at the process 189 in a time alignedmanner. The process 189 is identical to any of the novel beamformingmethods described above with respect to the two element broadsidearrays. Due to the axial symmetry about the sensitivity axis I, thispencil beam configuration produces a sensitivity beam with restrictedsensitivity in both the X and Y directions.

The inventive apparatus and method need not be used only for thecreation of narrow sensitivity beams. It can also be used to increasethe width of wide sensitivity beams, in other words to narrow the widthof the nulls between sensitivity lobes. Such operation is valuable in anumber of applications, for example in a type of beamformer systemcalled the generalized side lobe canceller (GSC). The best known GSC isthe Griffiths-Jim beamformer, originally proposed as a means to improvethe performance of radio-frequency antenna systems.

In the Griffiths-Jim beamformer, the signals from the array's broadsidearray sensor elements are combined by 1) a first method that captures asignal combining both the desired signal and the noise, and 2) secondmethods whose intention is to produce different signals that areversions of the noise only. The second method signals are produced by ablocking matrix that combines the sensor signals in ways that createnulls in the direction of the desired signal. Signals from the blockingmatrix are then modified by adaptive filters before being subtractedfrom the signal resulting from the first method so as to remove thenoise from this combined signal. The result is put out as the noisereduced final signal. Feedback from this noise reduced final signal isused to adapt the adaptive filter coefficients using a least meansquared (LMS) or other adaptation method in order to minimize theresidual noise in the noise reduced final signal.

Although this beamforming technique can be used with any number of arraysensors greater than one, for simplicity the two-sensor example will bediscussed here in reference to the application of the inventiveapparatus and method. FIG. 19(a) shows the structure of this type ofprior art two-element noise reduction system. The two-sensor broadsidearray 190 a is shown with the elements A and B, and it is assumed thatthe desired signal is arriving from a direction along the array axis LThe input signals from the two sensors are combined in the first methodby summing at circuit 191 (as in the classic delay-and-sum beamformer)the two sensor signals produced by the two array elements to generatethe combined signal labeled DS. This first signal has a beam shape thatvaries with frequency, being a FIG. 8 pattern with a maximum pointing inthe direction of arrival of the desired signal when the array elementsare separated by a distance of one-half wavelength, but with a nearlycircular beam pattern at frequencies well below the half wavelengthfrequency.

The same input signals are also combined in the second method bydifferencing the two signals at 192, to generate the second noise-onlysignal labeled NS. The differencing circuit 191 is the blocking matrixfor this two-element array example. The sensitivity pattern for thesecond signal has a FIG. 8 beam shape at all frequencies, but with anull pointing directly toward the desired signal and maximum sensitivityalong the orthogonal axis X. Thus, the signal DS contains both desiredsignal plus noise, while the second signal NS contains only noise.

The signal NS is then adaptively filtered by the filter 194, here shownas a digital finite impulse response filter (FIR). The adaptation ofthis filter is controlled by a least mean squared (LMS) circuit 195 thatattempts to minimize the noise power in the final output signal byadjusting the filter coefficients. The output from the filter circuit issubtracted from a delayed version of the combined signal DS at 196 toprovide the final noise reduced output signal. The time delay 193 isrequired to compensate for the time delay in the filter 194 so as totime align the combined signal DS with the noise-only signal created bythe filter 194, before subtraction at 196.

In this system, the width of the null in the beam pattern of the secondmethod signal NS determines the maximum noise reduction possible withthe Griffiths-Jim beamformer, with a narrower null producing a greaterpossible noise reduction. However in the prior art system shown in FIG.19(a), the width of the null is fixed and can not be varied, so themaximum amount of noise reduction is fixed.

If instead, as shown on FIG. 19(b), the blocking matrix's signaldifferencing circuit used in the Griffiths-Jim type beamformer ispreceded with the phase enhancement process, 198 of this invention, thenthe width of the null in the direction of the desired signal can bereduced in proportion to the amount of phase expansion, therebyachieving an increase of the noise reduction capabilities of theGriffiths-Jim beamformer.

Another noise reduction limitation of the prior art system shown in FIG.19(a) is a result of the frequency variation in beam shape for thesignal DS. This variation produces a different frequency responsecharacteristic for every angle of arrival. Thus, off-axis noise sourcesare “colored”, and the adaptive filter must re-adapt whenever there isrelative motion between the sensor array and the noise source orsources. During the time that the filter is adapting, the noise is notreduced, but is instead passed to the output.

By substituting the novel process of this invention for the first signalmethod 191 in FIG. 19(a), used to create the signal DS, the frequencyvariation can be eliminated, since the value of the parameter S can beadjusted to compensate for the frequency variation. This is shown inFIG. 19(b) by the circuit block 197. Elimination of the frequencyvariation of beam shape eliminates the frequency response variationcaused by the relative motion of off-axis noise signals, and therebyreduces or eliminates the re-adaptation time caused by relative motionbetween the sensor array and noise sources.

In wideband applications, the Griffiths-Jim beamformer and many otherGSC's will not operate with end-fire sensor array configurations. Thislimitation is due to the need to maintain, at all frequencies, a beampattern null in the direction of the desired source for the blockingmatrix signals. For example, in the system shown in FIG. 19(a), if thesensor array were to be configured as an end-fire array, as shown as 190b in FIG. 19(c), the signals NS and DS in FIG. 19(a) would beinterchanged, since the first method and second method beams would berotated by 90 degrees. However as described above, the beam patternformed by the summation circuit 191 only has a null when the sensorelements are spaced one-half wavelength apart. Since the one halfwavelength condition only occurs at a single frequency, such a systemonly operates correctly for frequencies at or near to that frequency.Because the circuit element 191 is producing the signal NS in thisconfiguration, at frequencies away from the one half wavelengthfrequency the null disappears and some of the desired signal “leaks”into the noise canceling adaptive filter. The result is that a part ofthe desired signal is undesirably removed from the output signalcreating distortion of the desired signal. Thus, it is possible tocreate an end-fire version of the conventional Griffiths-Jim beamformer,but only for very narrow bandwidth applications where the elementspacing can be made equal to one-half wavelength.

If instead, for the end-fire array configuration 190b of FIG. 19(c), thenoise signal NS is produced by a blocking matrix 192 consisting of thenovel phase enhancement process, then the variation of the noise sensingbeam pattern can be eliminated by frequency tapering of the values usedfor the sharpness parameter S. Thus, the null toward the desired signalcan be maintained constant over frequency, as is required for correctoperation of the adaptive noise reduction process of the GSC.

Similarly, the combined signal DS can be obtained in this end-fireconfiguration by application of the novel phase enhancement prior to avector signal difference circuit, as is shown at 199 in FIG. 19(c).However the system shown in FIG. 18(c) could be alternativelysubstituted for circuit block 199 in FIG. 19(c) in order to produce thecombined signal DS.

Most of the above described beamforming systems utilize additivebeamforming methods, where the phase-enhanced signals are summed toproduce the output signal. However there is another class of beamformersthat use only signal differences to create the beam patterns ofinterest. These beamformers are called subtractive beamformers, and thesimplest is the two-element end-fire array.

Phase compression, the reverse of phase expansion, can be usedbeneficially in subtractive beamforming array systems. For example,utilizing two omni-directional microphone elements, an acoustic end-firebeamformer is created when the rear element's signal is subtracted fromthe front element's signal. For example, in acoustic pickup sensorapplications, the resulting beam pattern is an end-fire FIG. 8, commonlycalled a noise canceling microphone system.

By compressing the electrical phase difference between the two inputsignals before subtraction, the beam pattern can be narrowed. In otherwords, the beam pattern is made desirably less sensitive to off-axisnoise pickup. In the configuration shown in FIG. 19(c), the combinedsignal DS may be produced by first phase enhancing the sensor signalsaccording to the innovative methods described above and thendifferencing those signals, as shown by circuit block 199 in FIG, 19(c).In this case, the phase enhancement is preferably a phase compression.

As an example of phase compression, in Equation 1, phase compression isaccomplished by using values for the parameter, S, of between 0 and 1,in other words, 0≦S<1. The curve for this particular signal phasecompression characteristic is shown in FIG. 6(a) as graph curve 61. Thiscurve results when the value of 2 is used for the sharpness parameter S.Many other phase compression functions and curves are possible, and itis contemplated that any such function or curve can be used within thescope of this invention without limitation. Similarly, as shown byEquation 4, there are corresponding attenuation functions and these arealso contemplated to be unlimited within the scope of this invention.

Alternative to the frequency domain processing described above, theprocess can be applied in the time domain, wherein for example the inputsignal, either analog or digitized, is passed through a bank of bandpassfrequency discrimination filters (either analog or digital asappropriate). The outputs of each of the frequency filters issubsequently processed, for example by using the Hilbert transform tocreate an analytic signal for each input signal channel. The analyticsignals are then used to calculate in real-time the instantaneous phaseand instantaneous phase difference as well as the instantaneous signalmagnitudes, using methods that are well known in the art. The phasedifference is then used to, for example, attenuate the signal magnitudesas a function of the phase difference, using any of the attenuationfunctions, or a look-up table, as described above, before the processedsignals are then combined to form a processed output signal by addingtogether the processed signals. Alternatively, the instantaneous signalelectrical phase difference can be enhanced using any of the enhancementfunctions, or a look-up table, before combining the signals to form theprocessed output signal by adding together the phase expanded signals.

Additionally, the novel signal matching method can be applied withinsuch time domain processing techniques, by reassigning the individualinstantaneous signal magnitudes to be the mathematical mean value of theindividual signal magnitudes.

In the time domain processing technique, it is often desirable to filterthe measured parameters or the processing modifications to reducespurious effects that create noise in the process. Such filtering iscontemplated to be within the scope of the invention. The above timedomain methods are exemplary modes of carrying out the invention and arenot intended to be limiting.

It will be appreciated that while mostly herein described in terms ofaudio signals from a pair of microphones arranged as a broadside array,the method and system of the invention are applicable to any number ofsensor elements of all types, arranged in one, two or three dimensions.Uses of microphones, or other sound sensor elements generally, can be invehicle cabins (telephones, command and control) for civilian andmilitary uses, PCs, tablet PCs, PDAs, appliances, conference telephones,microphone arrays (for example, on top of PC monitors), concerts,sporting events and other large gatherings. Further, the signalenhancement aspects of the invention are equally applicable to non-audiosignals, finding uses in virtually any wave energy system, for exampleultrasound and infrasound systems, sonar and sonar imaging, radar andradar imaging, X-rays and X-ray imaging, underwater warfare, echolocation, astronomy, medical applications, optical imaging, gravity wavedetection and location, infrared applications, and so forth.

The above are exemplary modes of carrying out the invention and are notintended to be limiting. It will be apparent to those of ordinary skillin the art that modifications thereto can be made without departure fromthe spirit and scope of the invention as set forth in the followingclaims.

1. A beamformer using a plurality of sensors each producing a sensorinput signal representable by an input vector having phase and magnitudecomponents, the beamformer comprising: a combining circuit for receivingthe sensor input signals and generating a combined signal therefrom; afirst differencing circuit for receiving the sensor input signals andgenerating a first difference signal therefrom; an adaptive filter forreceiving the difference signal and generating a filtered signaltherefrom; a second differencing circuit for receiving the filteredsignal and a delayed version of the combined signal and generating anoutput signal therefrom; and a phase difference enhancement circuit forenhancing a phase difference of input vectors representing sensor inputsignals from the plurality of sensors.
 2. The beamformer of claim 1,wherein the combined signal represents the sum of the sensor inputsignals.
 3. A beamformer using a plurality of sensors each producing asensor input signal representable by an input vector having phase andmagnitude components, the beamformer comprising: a combining circuit forreceiving the sensor input signals and generating a combined signaltherefrom; a first differencing circuit for receiving the sensor inputsignals and generating a first difference signal therefrom; an adaptivefilter for receiving the difference signal and generating a filteredsignal therefrom; a second differencing circuit for receiving thefiltered signal and a delayed version of the combined signal andgenerating an output signal therefrom; and a phase difference responsivecircuit for receiving at least one of the signals, and modifying thatsignal to produce a modified signal that is a function of the phasedifference of the input signals from the plurality of sensors.
 4. Abeamformer using a plurality of sensors each producing a sensor inputsignal representable by an input vector having phase and magnitudecomponents, the beamformer comprising: a processing circuit forreceiving the sensor input signals and generating a processed signaltherefrom, the processing circuit including a first phase differenceenhancement circuit for enhancing the phase difference of input vectorsrepresenting sensor input signals from the plurality of sensors; a firstdifferencing circuit for receiving the sensor input signals andgenerating a first difference signal therefrom; an adaptive filter forreceiving the difference signal and generating a filtered signaltherefrom; a second differencing circuit for receiving the filteredsignal and a delayed version of the processed signal and generating anoutput signal therefrom; and a second phase difference enhancementcircuit for enhancing a phase difference of input vectors representingsensor input signals from the plurality of sensors.
 5. The device ofclaim 4, wherein the first phase difference enhancement circuitincreases or decreases the input phase difference in afrequency-dependent manner.
 6. The device of claim 5, wherein increasingis effected using an expansion function.
 7. The device of claim 5,wherein increasing is effected using a look-up table.
 8. The device ofclaim 4, wherein the first phase difference enhancement circuit performsenhancement as a function of an adjustable sharpness parameter.
 9. Thedevice of claim 8, wherein the adjustable sharpness parameter is appliedmultiplicatively.
 10. The device of claim 8, wherein the adjustablesharpness parameter is inversely proportional to frequency such thatuniform sensitivity across the frequency spectrum is achieved.
 11. Thedevice of claim 8, wherein the adjustable sharpness parameter has one ofmultiple values depending on the sign of the phase difference betweenthe first input vector and the second input vector.
 12. The device ofclaim 4, wherein the sensors are arranged in an end-fire array.
 13. Thedevice of claim 4, wherein the sensors are arranged in a broadsidearray.